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MC33166 Schematic ( PDF Datasheet ) - Motorola Semiconductors

Teilenummer MC33166
Beschreibung POWER SWITCHING REGULATORS
Hersteller Motorola Semiconductors
Logo Motorola Semiconductors Logo 




Gesamt 16 Seiten
MC33166 Datasheet, Funktion
Order this document by MC34166/D
Power Switching Regulators
MC34166
MC33166
The MC34166, MC33166 series are high performance fixed frequency
power switching regulators that contain the primary functions required for
dc–to–dc converters. This series was specifically designed to be
incorporated in step–down and voltage–inverting configurations with a
minimum number of external components and can also be used cost
effectively in step–up applications.
These devices consist of an internal temperature compensated
reference, fixed frequency oscillator with on–chip timing components,
latching pulse width modulator for single pulse metering, high gain error
amplifier, and a high current output switch.
Protective features consist of cycle–by–cycle current limiting,
undervoltage lockout, and thermal shutdown. Also included is a low power
standby mode that reduces power supply current to 36 µA.
Output Switch Current in Excess of 3.0 A
Fixed Frequency Oscillator (72 kHz) with On–Chip Timing
Provides 5.05 V Output without External Resistor Divider
Precision 2% Reference
0% to 95% Output Duty Cycle
Cycle–by–Cycle Current Limiting
Undervoltage Lockout with Hysteresis
Internal Thermal Shutdown
Operation from 7.5 V to 40 V
Standby Mode Reduces Power Supply Current to 36 µA
Economical 5–Lead TO–220 Package with Two Optional Leadforms
Also Available in Surface Mount D2PAK Package
POWER SWITCHING
REGULATORS
SEMICONDUCTOR
TECHNICAL DATA
TH SUFFIX
PLASTIC PACKAGE
CASE 314A
1
5
1
5
TV SUFFIX
PLASTIC PACKAGE
CASE 314B
Heatsink surface connected to Pin 3.
T SUFFIX
PLASTIC PACKAGE
CASE 314D
1
5
Simplified Block Diagram
(Step Down Application)
ILIMIT
Vin
4
Oscillator
PWM
Thermal
S
Q
R
UVLO
Reference
2
EA
1
35
This device contains 143 active transistors.
L
VO
5.05 V/3.0 A
Pin 1. Voltage Feedback Input
2. Switch Output
3. Ground
4. Input Voltage/VCC
5. Compensation/Standby
D2T SUFFIX
PLASTIC PACKAGE
CASE 936A
1 (D2PAK)
5
Heatsink surface (shown as terminal 6
in case outline drawing) is connected to Pin 3.
ORDERING INFORMATION
Operating
Device Temperature Range Package
MC33166D2T
Surface Mount
MC33166T TA = – 40° to + 85°C Straight Lead
MC33166TH
Horiz. Mount
MC33166TV
Vertical Mount
MC34166D2T
Surface Mount
MC34166T
MC34166TH
TA = 0° to + 70°C Straight Lead
Horiz. Mount
MC34166TV
Vertical Mount
MOTOROLA ANALOG IC DEVICE DATA
© Motorola, Inc. 1996
Rev 4
1






MC33166 Datasheet, Funktion
MC34166 MC33166
INTRODUCTION
The MC34166, MC33166 series are monolithic power
switching regulators that are optimized for dc–to–dc converter
applications. These devices operate as fixed frequency,
voltage mode regulators containing all the active functions
required to directly implement step–down and
voltage–inverting converters with a minimum number of
external components. They can also be used cost effectively
in step–up converter applications. Potential markets include
automotive, computer, industrial, and cost sensitive consumer
products. A description of each section of the device is given
below with the representative block diagram shown in
Figure 13.
Oscillator
The oscillator frequency is internally programmed to
72 kHz by capacitor CT and a trimmed current source. The
charge to discharge ratio is controlled to yield a 95%
maximum duty cycle at the Switch Output. During the
discharge of CT, the oscillator generates an internal blanking
pulse that holds the inverting input of the AND gate high,
disabling the output switch transistor. The nominal oscillator
peak and valley thresholds are 4.1 V and 2.3 V respectively.
Pulse Width Modulator
The Pulse Width Modulator consists of a comparator with
the oscillator ramp voltage applied to the noninverting input,
while the error amplifier output is applied into the inverting
input. Output switch conduction is initiated when CT is
discharged to the oscillator valley voltage. As CT charges to
a voltage that exceeds the error amplifier output, the latch
resets, terminating output transistor conduction for the
duration of the oscillator ramp–up period. This PWM/Latch
combination prevents multiple output pulses during a given
oscillator clock cycle. Figures 6 and 14 illustrate the switch
output duty cycle versus the compensation voltage.
Current Sense
The MC34166 series utilizes cycle–by–cycle current
limiting as a means of protecting the output switch transistor
from overstress. Each on–cycle is treated as a separate
situation. Current limiting is implemented by monitoring the
output switch transistor current buildup during conduction, and
upon sensing an overcurrent condition, immediately turning off
the switch for the duration of the oscillator ramp–up period.
The collector current is converted to a voltage by an
internal trimmed resistor and compared against a reference
by the Current Sense comparator. When the current limit
threshold is reached, the comparator resets the PWM latch.
The current limit threshold is typically set at 4.3 A. Figure 9
illustrates switch output current limit threshold versus
temperature.
Error Amplifier and Reference
A high gain Error Amplifier is provided with access to the
inverting input and output. This amplifier features a typical dc
voltage gain of 80 dB, and a unity gain bandwidth of
600 kHz with 70 degrees of phase margin (Figure 3). The
noninverting input is biased to the internal 5.05 V reference
and is not pinned out. The reference has an accuracy of
± 2.0% at room temperature. To provide 5.0 V at the load, the
reference is programmed 50 mV above 5.0 V to compensate
for a 1.0% voltage drop in the cable and connector from the
converter output. If the converter design requires an output
voltage greater than 5.05 V, resistor R1 must be added to
form a divider network at the feedback input as shown in
Figures 13 and 18. The equation for determining the output
+ ǒ ) Ǔvoltage with the divider network is:
Vout
5.05
R2
R1
1
External loop compensation is required for converter
stability. A simple low–pass filter is formed by connecting a
resistor (R2) from the regulated output to the inverting input,
and a series resistor–capacitor (RF, CF) between Pins 1 and
5. The compensation network component values shown in
each of the applications circuits were selected to provide
stability over the tested operating conditions. The step–down
converter (Figure 18) is the easiest to compensate for
stability. The step–up (Figure 20) and voltage–inverting
(Figure 22) configurations operate as continuous conduction
flyback converters, and are more difficult to compensate. The
simplest way to optimize the compensation network is to
observe the response of the output voltage to a step load
change, while adjusting RF and CF for critical damping. The
final circuit should be verified for stability under four boundary
conditions. These conditions are minimum and maximum
input voltages, with minimum and maximum loads.
By clamping the voltage on the error amplifier output
(Pin 5) to less than 150 mV, the internal circuitry will be
placed into a low power standby mode, reducing the power
supply current to 36 µA with a 12 V supply voltage. Figure 10
illustrates the standby supply current versus supply voltage.
The Error Amplifier output has a 100 µA current source
pull–up that can be used to implement soft–start. Figure 17
shows the current source charging capacitor CSS through a
series diode. The diode disconnects CSS from the feedback
loop when the 1.0 M resistor charges it above the operating
range of Pin 5.
Switch Output
The output transistor is designed to switch a maximum of
40 V, with a minimum peak collector current of 3.3 A. When
configured for step–down or voltage–inverting applications,
as in Figures 18 and 22, the inductor will forward bias the
output rectifier when the switch turns off. Rectifiers with a
high forward voltage drop or long turn–on delay time should
not be used. If the emitter is allowed to go sufficiently
negative, collector current will flow, causing additional device
heating and reduced conversion efficiency. Figure 8 shows
that by clamping the emitter to 0.5 V, the collector current will
be in the range of 100 µA over temperature. A 1N5822 or
equivalent Schottky barrier rectifier is recommended to fulfill
these requirements.
Undervoltage Lockout
An Undervoltage Lockout comparator has been
incorporated to guarantee that the integrated circuit is fully
functional before the output stage is enabled. The internal
5.05 V reference is monitored by the comparator which
enables the output stage when VCC exceeds 5.9 V. To
prevent erratic output switching as the threshold is crossed,
0.9 V of hysteresis is provided.
6 MOTOROLA ANALOG IC DEVICE DATA

6 Page









MC33166 pdf, datenblatt
MC34166 MC33166
Figure 25. Negative Input/Positive Output Regulator
+
ILIMIT
4
Oscillator
PWM
Thermal
S
Q
R
+
UVLO
Reference
EA
Q1
2
+
1
22
0.01
1N5822
ǒ Ǔ+ )VO
5.05
R1
R2
0.7
R1
D1
Z1
L
MUR415
MTP
3055E
VO
+ 36 V/0.25 A
+
R1 1000
36 k
2N3906
3
Vin
–12 V
+
5 0.22 470 k
0.002
R2
5.1 k
1000
*Gate resistor RG, zener diode D3, and diode D4 are required only when Vin is greater than 20 V.
Test
Conditions
Results
Line Regulation
Vin = –10 V to – 20 V, IO = 0.25 A
250 mV = ± 0.35%
Load Regulation
Vin = –12 V, IO = 0.025 A to 0.25 A
790 mV = ±1.19%
Output Ripple
Vin = –12 V, IO = 0.25 A
80 mVpp
Efficiency
Vin = –12 V, IO = 0.25 A
79.2%
L = Coilcraft M1496–A or ELMACO CHK1050, 42 turns of #16 AWG on Magnetics Inc. 58350–A2 core.
Heatsink = AAVID Engineering Inc. 5903B or 5930B
Figure 26. Variable Motor Speed Control with EMF Feedback Sensing
+
Oscillator
PWM
ILIMIT
S
Q
R
UVLO
Vin
4 18 V
+
1000
2 1N5822
Thermal
+ Reference
EA
Brush
Motor
+
1
5.6 k 1.0 k
+
47
50 k
Faster
3 5 0.1 56 k
Test
Low Speed Line Regulation
High Speed Line Regulation
Conditions
Vin = 12 V to 24 V
Vin = 12 V to 24 V
Results
1760 RPM ±1%
3260 RPM ± 6%
12 MOTOROLA ANALOG IC DEVICE DATA

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